TDA8920BTH  by  Philips high efficiency (2X100W) D audio amplifier, IC use HSOP24 power package, with low power consumption.

The Class D audio amplifier has zero dead-switching, advanced current protection, smooth start work, there is no general amplifier boot (flapping – flapping) the impact of sound. Complete protection: over current, over voltage, over temperature protection, as a stereo single-ended amplifier (gain 30dB) or BTL mono amplifiers (gain 36dB), the most suitable for home audio, multimedia systems, car audio applications.

Good amplifier IC is a very important first and foremost, a good application circuit is also necessary requirements. The digital audio amplifier, and the emphasis is on the most critical PCB board layout techniques and technology, more than six months after the board changes and a series of optimization, the end achieved very good results, whether from the electrical properties of the test indicators, such as distortion degrees, SNR, dynamic range, etc., or from listening In effect, can be regarded as a comprehensive evaluation of very good class D amplifier.
This version supports the direct exchange of dual power, use special custom anti-saturation high current power inductors, 1% precision chip resistors (in the PCB bottom), U.S. DIODES rectifier, . Matching transformer output specification for the exchange of 2 * (16 ~ 27) V, power more than 200W .
PCB on the back of the circuit through the adjustment, you can easily work in dual channel or BTL bridge state.

Basic performance parameters:
D Class TDA8920BTH PHILIPS
Efficiency: 94%
Rated output power: 2X100W@4 Ohms
Frequency response: 20Hz to 50KHz
Power Supply Voltage: ± 12.5V to ± 30V
Recommended supply voltage: ± 26V
Maximum output current: 8A
The maximum operating temperature: 150 ºC
PCB board size: (70 × 86) mm

Shaped series of high-power, DC Inverter Regulated Power Supply (DC-DC), is a wide voltage input, multiple output voltage and high current, high-quality, inverter DC regulated switching power supply. For a variety of cars, locomotives, industrial control, car audio Such as: instruments, meters, motor, water pump, car amplifier, LCD display, camera, CD / DVD machines, solar LED lamps, displays and so on. Shaped series of high-power DC switching power supply can be DC voltage DC12V, converted into single and double groups 24V, 28V, 32V, 36V, 48V, 56V, 64V, 72V and so the output voltage.
Weight: 550g ~ 700g (grams)
Size: 165mm * 145mm × 60mm (mm)
Metal shell
Shaped series of high-power Dual DC Power Supply Selection:

Model : CAR SMPS  DC12V ± 24V
Car Power supply 380w classd.co.uk
Input voltage     DC11V-14V
Output voltage  ± 24V/8.0A
Ripple coefficient Milivolts <150mv
Output power Watt 380W

PCB Size: 130mm (length), 100mm (width), 57mm
High, including the PCB at the bottom of 8mm high copper pillar).
Efficiency greater than 90%, without additional heat sink and fan, cooling can be.
Oscillation frequency of 80KHz.

Over-current protection, SD enable pin (do temperature protection).
Ripple is less than 1%.
Instantaneous output current can reach 5-10 times the normal current, very suitable for high-power amplifier.
Shop sale of switching power supply modules and switching power supply board drives all
CE, UL, CCC, ROHS certification,

Output voltage can be customized according to user requirements, such as + /-20VDC,
+ /-30VDC, + /-40VDC —- + /-100VDC and so on.

Audio Amplifier Background The goal of audio amplifiers is to reproduce input audio signals at sound-producing output elements, with desired volume and power levels—faithfully, efficiently, and at low distortion. Audio frequencies range from about 20 Hz to 20 kHz, so the amplifier must have good frequency response over this range (less when driving a band-limited speaker, such as a woofer or a tweeter).
Power capabilities vary widely depending on the application, from milliwatts in headphones, to a few watts in TV or PC audio, to tens of watts for “mini” home stereos and automotive audio, to hundreds of watts and beyond for more powerful home and commercial sound systems—and to fill theaters or auditoriums with sound.
A straightforward analog implementation of an audio amplifier uses transistors in linear mode to create an output voltage that is a scaled copy of the input voltage. The forward voltage gain is usually high (at least 40 dB). If the forward gain is part of a feedback loop, the overall loop gain will also be high. Feedback is often used because high loop gain improves performance— suppressing distortion caused by nonlinearities in the forward path and reducing power supply noise by increasing the power-supply rejection (PSR).
The Class D Amplifier Advantage In a conventional transistor amplifier, the output stage contains transistors that supply the instantaneous continuous output current. The many possible implementations for audio systems include Classes A, AB, and B. Compared with Class D designs, the output-stage power dissipation is large in even the most efficient linear output stages. This difference gives Class D significant advantages in many applications because the lower power dissipation produces less heat, saves circuit board space and cost, and extends battery life in portable systems.
Linear Amplifiers, Class D Amplifiers, and Power Dissipation Linear-amplifier output stages are directly connected to the speaker (in some cases via capacitors). If bipolar junction transistors (BJTs) are used in the output stage, they generally operate in the linear mode, with large collector-emitter voltages.
The output stage could also be implemented with MOS transistors, as shown in Figure 1.

Power is dissipated in all linear output stages, because the process of generating VOUT unavoidably causes nonzero IDS and VDS in at least one output transistor. The amount of power dissipation strongly depends on the method used to bias the
output transistors.
The Class A topology uses one of the transistors as a dc current source, capable of supplying the maximum audio current required
by the speaker. Good sound quality is possible with the Class A output stage, but power dissipation is excessive because a large dc bias current usually flows in the output-stage transistors (where we do not want it), without being delivered to the speaker (where we do want it).
The Class B topology eliminates the dc bias current and dissipates significantly less power. Its output transistors are individually controlled in a push-pull manner, allowing the MH device to supply positive currents to the speaker, and ML to sink negative currents. This reduces output stage power dissipation, with only signal current conducted through the transistors.
The Class B circuit has inferior sound quality, however, due to nonlinear behavior (crossover distortion) when the output current passes through 0 and the transistors are changing between the on and off conditions.
Class AB, a hybrid compromise of Classes A and B, uses some dc bias current, but much less than a pure Class A design.
The small dc bias current is sufficient to prevent crossover distortion, enabling good sound quality. Power dissipation, although between Class A and Class B limits, is typically closer to Class B. Some control, similar to that of the Class B circuit, is needed to allow the Class AB circuit to supply or sink large output currents.
Unfortunately, even a well-designed class AB amplifier has significant power dissipation, because its midrange output voltages are generally far from either the positive or negative supply rails. The large drain-source voltage drops thus produce
significant IDS 3 VDS instantaneous power dissipation.
Thanks to a different topology (Figure 2), the Class D amplifier dissipates much less power than any of the above. Its output stage switches between the positive and negative power supplies so as to produce a train of voltage pulses. This waveform is benign for power dissipation, because the output transistors have zero current when not switching, and have low VDS when they are conducting current, thus giving smaller IDS 3 VDS.

Since most audio signals are not pulse trains, a modulator must be included to convert the audio input into pulses. The frequency content of the pulses includes both the desired audio signal and significant high-frequency energy related to the modulation process. A low-pass filter is often inserted between the output stage and the speaker to minimize electromagnetic interference (EMI) and avoid driving the speaker with too much high frequency energy.
The filter (Figure 3) needs to be lossless (or nearly so) in order to retain the power-dissipation advantage of the switching output stage. The filter normally uses capacitors and inductors, with the only intentionally dissipative element being the speaker.

Figure 4 compares ideal output-stage power dissipation (PDISS) for Class A and Class B amplifiers with measured dissipation for the AD1994 Class D amplifier, plotted against power delivered to the speaker (PLOAD), given an audio-frequency sine wave signal.
The power numbers are normalized to the power level, PLOAD max, at which the sine is clipped enough to cause 10% total harmonic
distortion (THD). The vertical line indicates the PLOAD at which clipping begins.
Significant differences in power dissipation are visible for a wide range of loads, especially at high and moderate values. At the onset of clipping, dissipation in the Class D output stage is about 2.5 times less than Class B, and 27 times less than Class A. Note that more power is consumed in the Class A output stage than is delivered to the speaker—a consequence of using the large dc bias current. Output-stage power efficiency, Eff, is defined as

At the onset of clipping, Eff = 25% for the Class A amplifier, 78.5% for the Class B amplifier, and 90% for the Class D amplifier (see Figure 5). These best-case values for Class A and Class B are the ones often cited in textbooks.

The differences in power dissipation and efficiency widen at moderate power levels. This is important for audio, because longterm average levels for loud music are much lower (by factors of five to 20, depending on the type of music) than the instantaneous peak levels, which approach PLOAD max. Thus, for audio amplifiers, [PLOAD = 0.1 3 PLOAD max] is a reasonable average power level at which to evaluate PDISS. At this level, the Class D output-stage dissipation is nine times less than Class B, and 107 times less than Class A. For an audio amplifier with 10-W PLOAD max, an average PLOAD of 1 W can be considered a realistic listening level. Under this condition, 282 mW is dissipated inside the Class D output stage, vs. 2.53 W for Class B and 30.2 W for Class A. In this case, the Class D efficiency is reduced to 78%—from 90% at higher power. But even 78% is much better than the Class B and Class A efficiencies—28% and 3%, respectively.

These differences have important consequences for system design. For power levels above 1 W, the excessive dissipation of linear output stages requires significant cooling measures to avoid unacceptable heating—typically by using large slabs of metal as heat sinks, or fans to blow air over the amplifier. If the amplifier is implemented as an integrated circuit, a bulky and expensive thermally enhanced package may be needed to facilitate heat transfer. These considerations are onerous in consumer products such as flat-screen TVs, where space is at a premium—or automotive audio, where the trend is toward cramming higher channel counts into a fixed space. For power levels below 1 W, wasted power can be more of a difficulty than heat generation. If powered from a battery, a linear output stage would drain battery charge faster than a Class D design. In the above example, the Class D output stage consumes 2.8 times less supply current than Class B and 23.6 times less than Class A—resulting in a big difference in the life of batteries used in products like cell phones, PDAs, and MP3 players.
For simplicity, the analysis thus far has focused exclusively on the amplifier output stages. However, when all sources of power dissipation in the amplifier system are considered, linear amplifiers can compare more favorably to Class D amplifiers at low output-power levels. The reason is that the power needed to generate and modulate the switching waveform can be significant at low levels. Thus, the system-wide quiescent dissipation of well-designed low-to-moderate-power Class AB amplifiers can make them competitive with Class D amplifiers. Class D power dissipation is unquestionably superior for the higher output power ranges, though.
Class D Amplifier Terminology, and Differential vs. Single-Ended Versions Figure 3 shows a differential implementation of the output transistors and LC filter in a Class D amplifier. This H-bridge has two half-bridge switching circuits that supply pulses of opposite polarity to the filter, which comprises two inductors, two capacitors, and the speaker. Each half-bridge contains two output transistors—a high-side transistor (MH) connected to the positive power supply, and a low-side transistor (ML) connected to the negative supply. The diagrams here show high-side pMOS transistors. High-side nMOS transistors are often used to reduce size and capacitance, but special gate-drive techniques are required to control them (Further Reading 1).
Full H-bridge circuits generally run from a single supply (VDD), with ground used for the negative supply terminal (VSS). For a
given VDD and VSS, the differential nature of the bridge means that it can deliver twice the output signal and four times the output power of single-ended implementations. Half-bridge circuits can be powered from bipolar power supplies or a single supply, but the single-supply version imposes a potentially harmful dc bias voltage, VDD/2, across the speaker, unless a blocking capacitor is added.
The power supply voltage buses of half-bridge circuits can be “pumped” beyond their nominal values by large inductor currents from the LC filter. The dV/dt of the pumping transient can be limited by adding large decoupling capacitors between VDD and VSS. Full-bridge circuits do not suffer from bus pumping, because inductor current flowing into one of the half-bridges flows out of the other one, creating a local current loop that minimally disturbs the power supplies.

Factors in Audio Class D Amplifier Design
The lower power dissipation provides a strong motivation to use Class D for audio applications, but there are important challenges for the designer. These include:
• Choice of output transistor size
• Output-stage protection
• Sound quality
• Modulation technique
• EMI
• LC filter design
• System cost

Choice of Output Transistor Size
The output transistor size is chosen to optimize power dissipation over a wide range of signal conditions. Ensuring that VDS stays
small when conducting large IDS requires the on resistance (RON) of the output transistors to be small (typically 0.1 V to 0.2 V). But this requires large transistors with significant gate capacitance (CG).
The gate-drive circuitry that switches the capacitance consumes power—CV 2f, where C is the capacitance, V is the voltage change
during charging, and f is the switching frequency. This “switching loss” becomes excessive if the capacitance or frequency is too high, so practical upper limits exist. The choice of transistor size is therefore a trade-off between minimizing IDS 3 VDS losses during
conduction vs. minimizing switching losses. Conductive losses will dominate power dissipation and efficiency at high output power levels, while dissipation is dominated by switching losses at low output levels. Power transistor manufacturers try to minimize the RON 3 CG product of their devices to reduce overall power dissipation in switching applications, and to provide flexibility in
the choice of switching frequency.

Protecting the Output Stage
The output stage must be protected from a number of potentially hazardous conditions:
Overheating: Class D’s output-stage power dissipation, though lower than that of linear amplifiers, can still reach levels that endanger the output transistors if the amplifier is forced to deliver very high power for a long time. To protect against dangerous
overheating, temperature-monitoring control circuitry is needed.
In simple protection schemes, the output stage is shut off when its temperature, as measured by an on-chip sensor, exceeds a thermalshutdown safety threshold, and is kept off until it cools down. The sensor can provide additional temperature information, aside from the simple binary indication about whether temperature has exceeded the shutdown threshold. By measuring temperature, the control circuitry can gradually reduce the volume level, reducing power dissipation and keeping temperature well within
limits—instead of forcing perceptible periods of silence during thermal-shutdown events.
Excessive current flow in the output transistors: The low on resistance of the output transistors is not a problem if the output stage and speaker terminals are properly connected, but enormous currents can result if these nodes are inadvertently short-circuited to one another, or to the positive or negative power supplies. If unchecked, such currents can damage the transistors or surrounding circuitry. Consequently, current-sensing outputtransistor protection circuitry is needed. In simple protection schemes, the output stage is shut off if the output currents exceed a safety threshold. In more sophisticated schemes, the current-sensor output is fed back into the amplifier—seeking to limit the output current to a maximum safe level, while allowing the amplifier to run continuously without shutting down. In these schemes, shutdown can be forced as a last resort if the attempted limiting proves ineffective. Effective current limiters can also keep the amplifier running safely in the presence of momentarily large transient currents due to speaker resonances.
Undervoltage: Most switching output stage circuits work well only if the positive power supply voltages are high enough. Problems result if there is an undervoltage condition, where the supplies are too low. This issue is commonly handled by an undervoltage lockout circuit, which permits the output stages to operate only if the power supply voltages are above an undervoltage-lockout threshold.
Output transistor turn-on timing: The MH and ML output
stage transistors (Figure 6) have very low on resistance. It is therefore important to avoid situations in which both MH and ML are on
simultaneously, as this would create a low-resistance path from VDD to VSS through the transistors and a large shoot-through current.
At best, the transistors will heat up and waste power; at worst, the transistors may be damaged. Break-before-make control of the transistors prevents the shoot-through condition by forcing both transistors off before turning one on. The time intervals in which
both transistors are off are called nonoverlap time or dead time.

Sound Quality
Several issues must be addressed to achieve good overall sound quality in Class D amplifiers.
Clicks and pops, which occur when the amplifier is turning on or off can be very annoying. Unfortunately, however, they are easy to introduce into a Class D amplifier unless careful attention is paid to modulator state, output-stage timing, and LC filter state
when the amplifier is muted or unmuted.

Signal-to-noise ratio (SNR):
To avoid audible hiss from the amplifier noise floor, SNR should typically exceed 90 dB in low-power amplifiers for portable applications, 100 dB for medium-power designs, and 110 dB for high-power designs. This is achievable for a wide variety of amplifier implementations, but individual noise sources must be tracked during amplifier design to ensure a satisfactory overall SNR.

Distortion mechanisms:
These include nonlinearities in the modulation technique or modulator implementation—and the dead time used in the output stage to solve the shoot-through current problem.
Information about the audio signal level is generally encoded in the widths of the Class D modulator output pulses. Adding dead time to prevent output stage shoot-through currents introduces a nonlinear timing error, which creates distortion at the speaker in proportion to the timing error in relation to the ideal pulse width.
The shortest dead time that avoids shoot-through is often best for minimizing distortion; see Further Reading 2 for a detailed design method to optimize distortion performance of switching output stages.

Other sources of distortion include: mismatch of rise and fall times in the output pulses, mismatch in the timing characteristics for the output transistor gate-drive circuits, and nonlinearities in the components of the LC low-pass filter.

Power-supply rejection (PSR): In the circuit of Figure 2, power-supply noise couples almost directly to the speaker with very little rejection. This occurs because the output-stage transistors connect the power supplies to the low-pass filter through a very low resistance. The filter rejects high-frequency noise, but is designed to pass all audio frequencies, including noise. See Further Reading 3 for a good description of the effect of power-supply noise in singleended and differential switching output-stage circuits. If neither distortion nor power-supply issues are addressed, it is difficult to achieve PSR better than 10 dB, or total harmonic distortion (THD) better than 0.1%. Even worse, the THD tends to be the bad-sounding high-order kind.

Fortunately, there are good solutions to these issues. Using feedback with high loop gain (as is done in many linear amplifier designs) helps a lot. Feedback from the LC filter input will greatly improve PSR and attenuate all non-LC filter distortion mechanisms. LC filter nonlinearities can be attenuated by including the speaker in the feedback loop. Audiophile-grade sound quality with PSR > 60 dB and THD < 0.01% is attainable in well-designed closed-loop Class D amplifiers. Feedback complicates the amplifier design, however, because loop stability must be addressed (a nontrivial consideration for high-order design). Also, continuous-time analog feedback is necessary to capture important information about pulse timing errors, so the control loop must include analog circuitry to process the feedback signal. In integrated-circuit amplifier implementations, this can add to the die cost.
To minimize IC cost, some vendors prefer to minimize or eliminate analog circuit content. Some products use a digital open-loop modulator, plus an analog-to-digital converter to sense power-supply variations—and adjust the modulator’s behavior to compensate, as proposed in Further Reading 3. This can improve PSR, but will not address any of the distortion problems. Other digital modulators attempt to precompensate for expected output stage timing errors, or correct for modulator nonidealities. This can at least partly address some distortion mechanisms, but not all.
Applications that tolerate fairly relaxed sound-quality requirements can be handled by these kinds of open-loop Class D amplifiers, but some form of feedback seems necessary for best audio quality.
Modulation Technique
Class D modulators can be implemented in many ways, supported by a large quantity of related research and intellectual property.
This article will only introduce fundamental concepts.
All Class D modulation techniques encode information about the audio signal into a stream of pulses. Generally, the pulse widths
are linked to the amplitude of the audio signal, and the spectrum of the pulses includes the desired audio signal plus undesired
(but unavoidable) high-frequency content. The total integrated high-frequency power in all schemes is roughly the same, since
the total power in the time-domain waveforms is similar, and by Parseval’s theorem, power in the time domain must equal power in
the frequency domain. However, the distribution of energy varies widely: in some schemes, there are high energy tones atop a low noise floor, while in other schemes, the energy is shaped so that tones are eliminated but the noise floor is higher. The most common modulation technique is pulse-width modulation (PWM). Conceptually, PWM compares the input
audio signal to a triangular or ramping waveform that runs at a fixed carrier frequency. This creates a stream of pulses at
the carrier frequency. Within each period of the carrier, the duty ratio of the PWM pulse is proportional to the amplitude of the audio signal. In the example of Figure 7, the audio input and triangular wave are both centered around 0 V, so that for 0 input, the duty ratio of the output pulses is 50%.
For large positive input, it is near 100%, and it is near 0% for large negative input. If the audio amplitude exceeds that of the triangle wave, full modulation occurs, where the pulse train stops switching, and the duty ratio within individual periods is either 0% or 100%.
PWM is attractive because it allows 100-dB or better audio-band SNR at PWM carrier frequencies of a few hundred kilohertz—low
enough to limit switching losses in the output stage. Also, many PWM modulators are stable up to nearly 100% modulation, in concept permitting high output power—up to the point of overloading. However, PWM has several problems: First, the PWM process inherently adds distortion in many implementations (Further Reading 4); next, harmonics of the PWM carrier frequency produce EMI within the AM radio band; and finally, PWM pulse widths become very small near full modulation.
This causes problems in most switching output-stage gate-driver circuits—with their limited drive capability, they cannot switch properly at the excessive speeds needed to reproduce short pulses with widths of a few nanoseconds. Consequently, full modulation is often unattainable in PWM-based amplifiers, limiting maximum achievable output power to something less than the theoretical maximum—which considers only power-supply voltage, transistor on resistance, and speaker impedance.
An alternative to PWM is pulse-density modulation (PDM), in which the number of pulses in a given time window is proportional to the average value of the input audio signal. Individual pulse widths cannot be arbitrary as in PWM, but are instead “quantized” to multiples of the modulator clock period. 1-bit sigma-delta modulation is a form of PDM.
Much of the high-frequency energy in sigma-delta is distributed over a wide range of frequencies—not concentrated in tones at multiples of a carrier frequency, as in PWM—providing sigmadelta modulation with a potential EMI advantage over PWM.
Energy still exists at images of the PDM sampling clock frequency; but with typical clock frequencies from 3 MHz to 6 MHz, the images are outside the audio frequency band—and are strongly attenuated by the LC low-pass filter.
Another advantage of sigma-delta is that the minimum pulse width is one sampling-clock period, even for signal conditions approaching full modulation. This eases gate-driver design and allows safe operation to theoretical full power. Nonetheless 1-bit sigma-delta modulation is not often used in Class D amplifiers (Further Reading 4) because conventional 1-bit modulators are only stable to 50% modulation. Also, at least 643 oversampling is needed to achieve sufficient audio-band SNR, so typical output data rates are at least 1 MHz and power efficiency is limited.
Recently, self-oscillating amplifiers have been developed, such as the one in Further Reading 5. This type of amplifier always includes a feedback loop, with properties of the loop determining the switching frequency of the modulator, instead of an externally provided clock. High-frequency energy is often more evenly distributed than in PWM. Excellent audio quality is possible, thanks to the feedback, but the loop is self-oscillating, so it’s difficult to synchronize with any other switching circuits, or to connect to digital audio sources without first converting the digital to analog.
The full-bridge circuit (Figure 3) can use “3-state” modulation to reduce differential EMI. With conventional differential operation, the output polarity of Half-bridge A must be opposite to that of Half-bridge B. Only two differential operating states exist: Output A high with Output B low; and A low with B high. Two additional common-mode states exist, however, in which both half-bridge outputs are the same polarity (both high or both low).
One of these common-mode states can be used in conjunction with the differential states to produce 3-state modulation where the
differential input to the LC filter can be positive, 0, or negative. The 0 state can be used to represent low power levels, instead of switching between the positive and negative state as in a 2-state scheme. Very little differential activity occurs in the LC filter during the 0 state, reducing differential EMI, although actually increasing common-mode EMI. The differential benefit only applies at low power levels, because the positive and negative states must still be used to deliver significant power to the speaker. The varying common-mode voltage level in 3-state modulation schemes presents a design challenge for closed-loop amplifiers.

Taming EMI
The high-frequency components of Class D amplifier outputs merit serious consideration. If not properly understood and managed, these components can generate large amounts of EMI and disrupt operation of other equipment.
Two kinds of EMI are of concern: signals that are radiated into space and those that are conducted via speaker- and powersupply
wires. The Class D modulation scheme determines a baseline spectrum of the components of conducted and radiated EMI. However, some board-level design techniques can be used to reduce the EMI emitted by a Class D amplifier, despite its baseline spectrum.
A useful principle is to minimize the area of loops that carry high-frequency currents, since strength of associated EMI is related to loop area and the proximity of loops to other circuits.
For example, the entire LC filter (including the speaker wiring) should be laid out as compactly as possible, and kept close to the amplifier. Traces for current drive and return paths should be kept together to minimize loop areas (using twisted pairs for the speaker wires is helpful). Another place to focus is on the large charge transients that occur while switching the gate capacitance of the output-stage transistors. Generally this charge comes from a reservoir capacitance, forming a current loop containing both capacitances. The EMI impact of transients in this loop can be diminished by minimizing the loop area, which means placing the reservoir capacitance as closely as possible to the transistor(s) it charges.
It is sometimes helpful to insert RF chokes in series with the power supplies for the amplifier. Properly placed, they can confine high-frequency transient currents to local loops near the amplifier, instead of being conducted for long distances down the power supply wires.
If gate-drive nonoverlap time is very long, inductive currents from the speaker or LC filter can forward-bias parasitic diodes at the terminals of the output-stage transistors. When the nonoverlap time ends, the bias on the diode is changed from forward to reverse. Large reverse-recovery current spikes can flow before the diode fully turns off, creating a troublesome source of EMI. This problem can be minimized by keeping the nonoverlap time very short (also recommended to minimize distortion of the audio). If the reverse-recovery behavior is still unacceptable, Schottky diodes can be paralleled with the transistor’s parasitic diodes, in order to divert the currents and prevent the parasitic diode from ever turning on. This helps because the metal-semiconductor junctions of Schottky diodes are intrinsically immune to reverse-recovery effects. LC filters with toroidal inductor cores can minimize stray field lines resulting from amplifier currents. The radiation from the cheaper drum cores can be reduced by shielding, a good compromise between cost and EMI performance—if care is taken to ensure that the shielding doesn’t unacceptably degrade inductor linearity and sound quality at the speaker.

LC Filter Design
To save on cost and board space, most LC filters for Class D amplifiers are second-order, low-pass designs. Figure 3 depicts the differential version of a second-order LC filter. The speaker serves to damp the circuit’s inherent resonance. Although the speaker
impedance is sometimes approximated as a simple resistance, the actual impedance is more complex and may include significant reactive components. For best results in filter design, one should always seek to use an accurate speaker model.
A common filter design choice is to aim for the lowest bandwidth for which droop in the filter response at the highest audio frequency of interest is minimized. A typical filter has 40-kHz Butterworth response (to achieve a maximally flat pass band), if droop of less than 1 dB is desired for frequencies up to 20 kHz. The nominal component values in the table give approximate Butterworth response for common speaker impedances and standard L and C values:

If the design does not include feedback from the speaker, THD at the speaker will be sensitive to linearity of the LC filter components.
Inductor Design Factors: Important factors in designing or selecting the inductor include the core’s current rating and shape, and the winding resistance. Current rating: The core that is chosen should have a current rating above the highest expected amplifier current. The reason is that many inductor cores will magnetically saturate if current exceeds the current-rating threshold and flux density becomes too high—resulting in unwanted drastic reduction of inductance.
The inductance is formed by wrapping a wire around the core. If there are many turns, the resistance associated with the total wire length is significant. Since this resistance is in series between the half-bridge and the speaker, some of the output power will be dissipated in it. If the resistance is too high, use thicker wire or change the core to a different material that requires fewer turns of wire to give the desired inductance.
Finally, it should not be forgotten that the form of inductor used can affect EMI, as noted above.

System Cost
What are the important factors in the overall cost of an audio system that uses Class D amplifiers? How can we minimize the cost?
The active components of the Class D amplifier are the switching output stage and modulator. This circuitry can be built for roughly the same cost as an analog linear amplifier. The real trade-offs occur when considering other components of the system.
The lower dissipation of Class D saves the cost (and space) of cooling apparatus like heat sinks or fans. A Class D integratedcircuitamplifier may be able to use a smaller and cheaper package than is possible for the linear one. When driven from a digital audio source, analog linear amplifiers require D/A converters (DACs) to convert the audio into analog form. This is also true for analog-input Class D amplifiers, but digital-input types effectively integrate the DAC function.
On the other hand, the principal cost disadvantage of Class D is the LC filter. The components—especially the inductors—occupy board space and add expense. In high-power amplifiers, the overall system cost is still competitive, because LC filter cost is offset by large savings in cooling apparatus. But in cost-sensitive, low-power applications, the inductor expense becomes onerous. In extreme cases, such as cheap amplifiers for cell phones, an amplifier IC can be cheaper than the total LC filter cost. Also, even if the monetary cost is ignored, the board space occupied by the LC filter can be an issue in small form-factor applications.

To address these concerns, the LC filter is sometimes eliminated entirely, to create a filterless amplifier. This saves cost and space, though losing the benefit of low-pass filtering. Without the filter, EMI and high-frequency power dissipation can increase
unacceptably—unless the speaker is inductive and kept very close to the amplifier, current-loop areas are minimal, and power levels are kept low. Though often possible in portable applications like cell phones, it is not feasible for higher-power systems such as home stereos.
Another approach is to minimize the number of LC filter components requi red per audio channel. This can be accomplished by using single-ended half-br idge output stages, which require half the number of Ls and Cs needed for differential, full-bridge circuits. But if the half-bridge requires bipolar power supplies, the expense associated with generating the negative supply may be prohibitive, unless a negative supply is already present for some other purpose—or the amplifier has enough audio channels, to amortize the cost of the negative supply. Alternatively, the half-bridge could be powered from a single supply, but this reduces output power and often requires a large dc blocking capacitor.

Analog Devices Class D Amplifiers
All of the design challenges just discussed can add up to a rather demanding project. To save time for the designer, Analog Devices offers a variety of Class D amplifier integrated circuits,1 incorporating programmable-gain amplifiers, modulators, and
power output stages. To simplify evaluation, demonstration boards are available for each amplifier type to simplify evaluation.
The PCB layout and bill-of-materials for each of these boards serve as a workable reference design, helping customers quickly design working, cost-effective audio systems without having to “reinvent the wheel” to solve the major Class D amplifier design challenges.
Consider, for example, the AD1990,2 AD1992,3 and AD1994,4— a family of dual-amplifier ICs, targeted at moderate-power stereo or mono applications requiring two channels with output-per-channel of up to 5-, 10-, and 25 W, respectively.
Here are some properties of these ICs:
The AD1994 Class D audio power amplifier combines two programmable-gain amplifiers, two sigma-delta modulators, and two power-output stages to drive full H-bridge-tied loads in home theater-, automotive-, and PC audio applications.
It generates switching waveforms that can drive stereo speakers at up to 25 W per speaker, or a single speaker to 50 W monophonic, with 90% efficiency. Its single-ended inputs are applied to a programmable-gain amplifier (PGA) with gains settable to 0-, 6-, 12-, and 18 dB, to handle lowlevel signals.
The device has integrated protection against output-stage hazards of overheating, overcurrent, and shoot-through current. There are minimal clicks and pops associated with muting, thanks to special timing control, soft start, and dc offset calibration. Specifications include 0.001% THD, 105-dB dynamic range, and >60 dB PSR, using continuoustime analog feedback from the switching output stage and optimized output stage gate drive. Its 1-bit sigma-delta modulator is especially enhanced for the Class D application to achieve average data frequency of 500 kHz, with high loop gain to 90% modulation, and stability to full modulation. A standalone modulator mode allows it to drive external FETs for higher output power.
It uses a 5-V supply for the PGA, modulator, and digital logic, and a high-voltage supply from 8 V to 20 V for the
switching output stage. The associated reference design meets FCC Class B EMI requirements. When driving 6 V
loads with 5-V and 12-V supplies, the AD1994 dissipates 487 mW quiescently, 710 mW at the 2 3 1-W output level, and 0.27 mW in power-down mode. Available in a 64-lead LFCSP package, it is specified from –408C to +858C More technical information about Class D amplifiers—including implementations with Blackfin processors—can be found in the Further Reading section.

The red signal is Vin and the blue signal is the modulation signal Vm. The pulse width modulated signal Vd can be expressed as:
Vd = “1″ if Vin > Vm
Now we have a discreet or digital signal which can only be either “0″ or “1″. The signal has a fundamental frequency equal to that of the modulation frequency fm (the frequency of the modulation signal Vm) but will also contain the input signal and a band of frequency components around the modulation frequency. If the modulation frequency is much higher than that of the input signal Vin

Introduction

Most audio system design engineers are well aware of the power-efficiency advantages of Class D amplifiers over linear audio-amplifier classes such as Class A, B, and AB. In linear amplifiers such as Class AB, significant amounts of power are lost due to biasing elements and the linear operation of the output transistors. Because the transistors of a Class D amplifier are simply used as switches to steer current through the load, minimal power is lost due to the output stage. Any power losses associated with a Class D amplifier are primarily attributed to output transistor on-resistances, switching losses, and quiescent current overhead. Most power lost in an amplifier is dissipated as heat. Because heatsink requirements can be greatly reduced or eliminated in Class D amplifiers, they are ideal for compact high-power applications.

In the past, the power-efficiency advantage of classical PWM-based Class D amplifiers has been overshadowed by external filter component cost, EMI/EMC compliance, and poor THD+N performance when compared to linear amplifiers. However, most current-generation Class D amplifiers utilize advanced modulation and feedback techniques to mitigate these issues.

The Basics of Class D Amplifiers

While there are a variety of modulator topologies used in modern Class D amplifiers, the most basic topology utilizes pulse-width modulation (PWM) with a triangle-wave (or sawtooth) oscillator. Figure 1 shows a simplified block diagram of a PWM-based, half-bridge Class D amplifier. It consists of a pulse-width modulator, two output MOSFETs, and an external lowpass filter (LF and CF) to recover the amplified audio signal. As shown in the figure, the p-channel and n-channel MOSFETs operate as current-steering switches by alternately connecting the output node to VDD and ground. Because the output transistors switch the output to either VDD or ground, the resulting output of a Class D amplifier is a high-frequency square wave. The switching frequency (fSW) for most Class D amplifiers is typically between 250kHz to 1.5MHz. The output square wave is pulse-width modulated by the input audio signal. PWM is accomplished by comparing the input audio signal to an internally generated triangle-wave (or sawtooth) oscillator. This type of modulation is also often referred to as “natural sampling” where the triangle-wave oscillator acts as the sampling clock. The resulting duty cycle of the square wave is proportional to the level of the input signal. When no input signal is present, the duty cycle of the output waveform is equal to 50%. Figure 2 illustrates the resulting PWM output waveform due to the varying input-signal level.

Figure 1. This simplified functional block diagram illustrates a basic half-bridge Class D amplifier.

Figure 2. The output-signal pulse widths vary proportionally with the input-signal magnitude.

In order to extract the amplified audio signal from this PWM waveform, the output of the Class D amplifier is fed to a lowpass filter. The LC lowpass filter shown in Figure 1 acts as a passive integrator (assuming the cutoff frequency of the filter is at least an order of magnitude lower than the switching frequency of the output stage) whose output is equal to the average value of the square wave. Additionally, the lowpass filter prevents high-frequency switching energy from being dissipated in the resistive load. Assume that the filtered output voltage (VO_AVG) and current (IAVG) remain constant during a single switching period. This assumption is fairly accurate because fSW is much greater than the highest input audio frequency. Therefore, the relationship between the duty cycle and resulting filtered output voltage can be derived using a simple time-domain analysis of the inductor voltage and current.

The instantaneous current flowing through the inductor is:

where VL(t) is the instantaneous voltage across the inductor using the sign convention shown in Figure 1.

Because the average current (IAVG) flowing into the load is assumed constant over one switching period, the inductor current at the beginning of the switching period (TSW) must be equal to the inductor current at the end of the switching period, as shown in Figure 3.

In mathematical terms, this means that:

Figure 3. Filter inductor current and voltage waveforms are shown for a basic half-bridge Class D amplifier.

Equation 2 shows that the integral of the inductor voltage over one switching period must be equal to 0. Using equation 2 and examining the VL(t) waveform shown in Figure 3, it is clear that the absolute values of the areas (AON and AOFF) must be equal to each other in order for equation 2 to be true. With this information, we can now derive an expression for the filtered output voltage in terms of the duty ratio of the switching waveform:

Substituting equations 4 and 5 into equation 3 gives the new equation:

Finally, solving for VO gives:

where D is the duty ratio of the output-switching waveform.

Using Feedback to Improve Performance

Many Class D amplifiers utilize negative feedback from the PWM output back to the input of the device. A closed-loop approach not only improves the linearity of the device, but also allows the device to have power-supply rejection. This contrasts with an open-loop amplifier, which inherently has minimal (if any) supply rejection. Because the output waveform is sensed and fed back to the input of the amplifier in a closed-loop topology, deviations in the supply rail are detected at the output and corrected by the control loop. The advantages of a closed-loop design come at the price of possible stability issues, as is the case with all systems utilizing feedback. Therefore, the control loop must be carefully designed and compensated to ensure stability under all operating conditions.

Typical Class D amplifiers operate with a noise-shaping type of feedback loop, which greatly reduces in-band noise due to the nonlinearities of the pulse-width modulator, output stage, and supply-voltage deviations. This topology is similar to the noise shaping used in sigma-delta modulators. To illustrate this noise-shaping function, Figure 4 shows a simplified block diagram of a 1st-order noise shaper. The feedback network typically consists of a resistive-divider network but, for simplicity, the example shown in Figure 4 uses a feedback ratio of 1. Also, the transfer function for the integrator has been simplified to equal 1/s because the gain of an ideal integrator is inversely proportional to frequency. It is also assumed that the PWM block has a unity-gain and zero-phase-shift contribution to the control loop. Using basic control-block analysis, the following expression can be derived for the output:

Figure 4. A control loop with 1st-order noise shaping for a Class D amplifier pushes most noise out of band.

Equation 8 shows that the noise term, En(s), is multiplied by a highpass filter function (noise-transfer function) while the input term, VIN(s), is multiplied by a lowpass filter function (signal-transfer function). The noise-transfer function’s highpass filter response shapes the noise of the Class D amplifier. If the cutoff frequency of the output filter is selected properly, most of the noise is pushed out of band (Figure 4). While the preceding example dealt with a 1st-order noise shaper, many modern Class D amplifiers utilize multi-order noise-shaping topologies to further optimize linearity and power-supply rejection.

Class-D Topologies—Half Bridge vs. Full Bridge

Many Class D amplifiers are also implemented using a full-bridge output stage. A full bridge uses two half-bridge stages to drive the load differentially. This type of load connection is often referred to as a bridge-tied load (BTL). As shown in Figure 5, the full-bridge configuration operates by alternating the conduction path through the load. This allows bidirectional current to flow through the load without the need of a negative supply or a DC-blocking capacitor.

Figure 5. A traditional full-bridge Class D output stage uses two half-bridge stages to drive the load differentially.

Figure 6 illustrates the output waveforms of traditional BTL, PWM-based, Class D amplifiers. In Figure 6, the output waveforms are complements of each other, which produce a differential PWM signal across the load. As with the half-bridge topology, an external LC filter is needed at the output to extract the low-frequency audio signals and prevent high-frequency energy from being dissipated in the load.

Figure 6. Traditional full-bridge Class D output waveforms complement each other, thus creating a differential PWM signal across the load.

A full-bridge Class D amplifier shares the same advantages of a Class AB BTL amplifier, but adds high power efficiency. The first advantage of BTL amplifiers is that they do not require DC-blocking capacitors on the outputs when operating from a single supply. The same is not true for a half-bridge amplifier as its output swings between VDD and ground and idles at 50% duty cycle. This means that its output has a DC offset equal to VDD/2. With a full-bridge amplifier, this offset appears on each side of the load, which means that zero DC current flows at the output. The second advantage they share is that they can achieve twice the output signal swing when compared to a half-bridge amplifier with the same supply voltage because the load is driven differentially. This results in a theoretical 4x increase in maximum output power over a half-bridge amplifier operating from the same supply.

A full-bridge Class D amplifier, however, requires twice as many MOSFET switches as a half-bridge topology. Some consider this to be a disadvantage, because more switches typically mean more conduction and switching losses. However, this generally is only true with high-output power amplifiers (> 10W) due to the higher output currents and supply voltages involved. For this reason, half-bridge amplifiers are typically used for high-power applications for their slight efficiency advantage. Most high-power full-bridge amplifiers exhibit power efficiencies in the range of 80% to 88% with 8Ω loads. However, half-bridge amplifiers like the MAX9742 achieve power efficiencies greater than 90% while delivering more than 14W per channel into 8Ω.

Eliminating the Output Filter—Filterless Modulation

One of the major drawbacks of traditional Class D amplifiers has been the need for an external LC filter. This need not only increases a solution’s cost and board space requirements, but also introduces the possibility of additional distortion due to filter component nonlinearities. Fortunately, many modern Class D amplifiers utilize advanced “filterless” modulation schemes to eliminate, or at least minimize, external filter requirements.

Figure 7 shows a simplified functional diagram of the MAX9700 filterless modulator topology. Unlike the traditional PWM BTL amplifier, each half bridge has its own dedicated comparator, which allows each output to be controlled independently. The modulator is driven with a differential audio signal and a high-frequency sawtooth waveform. When both comparator outputs are low, each output of the Class D amplifier is high. At the same time, the output of the NOR gate goes high, but is delayed by the RC circuit formed by RON and CON. Once the delayed output of the NOR gate exceeds a specified threshold, switches SW1 and SW2 close. This causes OUT+ and OUT- to go low and remain as such until the next sampling period begins. This scheme causes both outputs to be on for a minimum amount of time (tON(MIN)), which is set by the values of RON and CON. As shown in Figure 8, with zero input, the outputs are in phase with pulse widths equal to tON(MIN). As the audio input signals increase or decrease, one comparator trips before the other. This behavior, along with the minimum on-time circuitry, causes one output to vary its pulse width while the other output pulse width remains at tON(MIN) (Figure 8). This means that the average value of each output contains a half-wave rectified version of the output audio signal. Taking the difference of the average values of the outputs yields the complete output audio waveform.

Figure 7. This simplified functional diagram shows the MAX9700's filterless Class D modulator topography.

Figure 8. The input and output waveforms are shown for the MAX9700's filterless modulator topography.

Because the MAX9700’s outputs idle with in-phase signals, there is no differential voltage applied across the load, thereby minimizing quiescent power consumption without the need for an external filter. Rather than depend on an external LC filter to extract the audio signal from the output, Maxim’s filterless Class D amplifiers rely on the inherent inductance of the speaker load and the human ear to recover the audio signal. The speaker resistance (RE) and inductance (LE) form a 1st-order lowpass filter which has a cutoff frequency equal to:

With most speakers, this 1st-order rolloff is enough to recover the audio signal and prevent excessive amounts of high-frequency switching energy from being dissipated in the speaker resistance. Even if residual switching energy results in speaker movement, these frequencies are inaudible to the human ear and will not adversely affect the listening experience. When using filterless Class D amplifiers, the speaker load should remain inductive at the amplifier’s switching frequency to achieve maximum output-power capabilities.
Minimizing EMI with Spread-Spectrum Modulation
One disadvantage of filterless operation is the possibility of radiated EMI from the speaker cables. Because the Class D amplifier output waveforms are high-frequency square waves with fast-moving transition edges, the output spectrum contains a large amount of spectral energy at the switching frequency and integer multiples of the switching frequency. Without an external output filter located within close proximity of the device, this high-frequency energy can be radiated by the speaker cables. Maxim’s filterless Class D amplifiers help mitigate possible EMI problems through a modulation scheme known as spread-spectrum modulation.

Spread-spectrum modulation is accomplished by dithering or randomizing the switching frequency of the Class D amplifier. The switching frequency is typically varied up to ±10% of the nominal switching frequency. While the period of the switching waveform is varied randomly cycle-to-cycle, the duty cycle is not affected, thereby preserving the audio content of the switching waveform. Figures 9a and 9b show the wideband output spectrum of the MAX9700 to illustrate the effects of spread-spectrum modulation. Rather than having the spectral energy concentrated at the switching frequency and its harmonics, spread-spectrum modulation effectively spreads out the spectral energy of the output signal. In other words, the total amount of energy present in the output spectrum remains the same, but the total energy is redistributed over a wider bandwidth. This reduces the high-frequency energy peaks at the outputs, therefore minimizing the chances of EMI being radiated from the speaker cables. While it is possible that some spectral noise may redistribute into the audio band with spread-spectrum modulation, this noise is suppressed by the noise-shaping function of the feedback loop.

Figure 9a. The wideband output spectrum is shown for the MAX9700 using a fixed switching frequency.

Figure 9b. Spread-spectrum modulation redistributes the spectral energy of the MAX9700 over a wider bandwidth.

Many of Maxim’s filterless Class D amplifiers also allow the switching frequency to be synchronized to an external clock signal. This allows the user to manually set the switching frequency of the amplifier to a less-sensitive frequency range.

While spread-spectrum modulation significantly improves EMI performance of filterless Class D amplifiers, there is typically a practical limit on the length of the speaker cables that can be used before the device begins to fail FCC or CE radiated-emissions regulations. If a device fails radiated-emissions tests due to long speaker cables, an external output filter may be needed to provide additional attenuation of the high-frequency components of the output waveform. In many applications with moderate speaker cable lengths, ferrite bead/capacitor filters on the outputs will suffice. EMI performance is also very layout sensitive, so proper PCB-layout guidelines should be strictly followed to guarantee compliance with applicable FCC and CE regulations.

Conclusion
Recent advancements in Class D modulation techniques have allowed Class D amplifiers to flourish in applications where linear amplifiers once dominated. Modern Class D amplifiers include all of the advantages of Class AB amplifiers (i.e., good linearity and minimal board-space requirements) with the added bonus of high power efficiency. Currently, there are a wide variety of Class D amplifiers available, thus making them suitable for numerous applications. These applications range from low-power portable applications (e.g., cell phones, notebooks) in which battery life, board-space requirements, and EMI compliance are of utmost importance, to high-power applications (e.g., automotive sound systems or flat-panel displays) where minimizing heatsinking requirements and heat generation is vital. Having a fundamental understanding of Class D amplifiers and their recent technological advances will aid designers in selecting the correct amplifier for their application and allow them to successfully weigh the advantages and disadvantages of specific features.

is an electronic amplifier where all power devices (usually MOSFETs) are operated as binary switches. They are either fully on or fully off. Ideally, zero time is spent transitioning between those two states. Theoretical power efficiency of class D amplifiers is 100%. That is to say, all of the power supplied to it is delivered to the load, none is turned to heat. This is because a switch in its on state will conduct all current but has no voltage across it, hence no heat is dissipated. And when it is off, it will have the full supply voltage standing across it, but no current flows through it. Again, no heat is dissipated. Real-life power MOSFETs are not ideal switches, but practical efficiencies well over 90% are common.
By contrast, linear AB-class amplifiers are always operated with both current flowing through and voltage standing across the power devices. An ideal class B amplifier has a theoretical maximum efficiency of 78%.
The output of a class D amplifier is a square wave. Low pass LC-filtering smoothes the pulses out and restores the signal shape on the load.
The term “class-D” is sometimes misunderstood as meaning a “digital” amplifier. While many class-D amps are indeed digital, the quantization of the output signal at the power stage can be controlled by either an analog signal or a digital signal. Only in the latter case would an amplifier be using fully digital amplification.

Signal modulation

Output stages such as those used in pulse generators are examples of class D amplifiers. However, the term mostly applies to devices intended to reproduce signals with a bandwidth well below the switching frequency. These amplifiers use pulse-width modulation (PWM), pulse density modulation (sometimes referred to as pulse frequency modulation), or more advanced forms of modulation such as delta-sigma modulation[1] or sliding mode control.

The input signal is converted to a sequence of pulses whose average value is directly proportional to the instantaneous value of the signal at that time. The frequency of the pulses is typically ten or more times the highest frequency of interest in the input signal. The final switching output consists of a train of pulses whose width is a function of the amplitude & frequency of the signal being amplified, and hence these amplifiers are also called PWM amplifiers. The output contains, in addition to the required amplified signal, unwanted spectral components (i.e. the pulse frequency and its harmonics) that must be removed by a passive filter. The filter is usually made with (theoretically) lossless components like inductors and capacitors in order to maintain efficiency.

A PWM amplifier operates similarly to a switched-mode power supply (SMPS), except that a PWM amplifier is feeding a varying audio signal voltage into a relatively fixed load, while an SMPS feeds a fixed voltage into a varying load. A switching amplifier must not be confused with an amplifier that uses an SMPS. A switching amplifier may use any type of power supply but the amplifier itself uses switching of output devices though to achieve amplification.

One way to create the PWM signal is to use a high speed comparator (“C” in the block-diagram above) that compares a high frequency triangular wave and the audio input and generates a series of pulses such that the width of the pulses corresponds to the amplitude and frequency of the audio signal. The comparator then drives a switching controller which in turn drives a high-power switch (usually made of MOSFETs) which generates a high-power replica of the comparator’s PWM signal.

This PWM output is fed to a low-pass filter which removes the high-frequency switching components of the PWM signal to recover the audio information and feeds it to a loudspeaker. A suitably high switching frequency (or triangular waveform) is mandatory in order to obtain reasonably good frequency response and low distortion. Most class-D amplifiers use switching frequencies greater than 100 kHz. These high frequencies require most of the components in the amplifier to be capable of high speed operation.

Another way to create the PWM signal is adopted when a SPDIF signal or other form of digital feed is available. The digital signal is fed to a DSP that uses software to create the PWM signal. This drives the MOSFETs through a suitable gate driver chip.

Two significant design challenges for MOSFET driver circuits in class-D amplifiers are keeping dead times and linear mode operation as short as possible. “Dead time” is the period during a switching transition when both output MOSFETs are driven into Cut-Off Mode and both are “off”. Dead times need to be as short as possible to maintain an accurate low-distortion output signal, but dead times that are too short cause the MOSFET that is switching on to start conducting before the MOSFET that is switching off has stopped conducting. The MOSFETs effectively short the output power supply through themselves, a condition known as “shoot-through”. Meanwhile, the MOSFET drivers also need to drive the MOSFETs between switching states as fast as possible to minimize the amount of time a MOSFET is in Linear Mode, the state between Cut-Off Mode and Saturation Mode where the MOSFET is neither fully on nor fully off and conducts current with a significant resistance, creating significant heat. Driver failures that allow shoot-through and/or too much linear mode operation result in excessive losses and sometimes catastrophic failure of the MOSFETs.

Error Control

The actual output of the amplifier is not just dependent on the content of the modulated PWM signal. The power supply voltage directly amplitude-modulates the output voltage, dead time errors make the output impedance non-linear and the output filter has a strongly load-dependent frequency response. An effective way to combat errors, regardless of their source, is negative feedback. A feedback loop including the output stage can be made using a simple integrator. To include the output filter, a PID controller is used, sometimes with additional integrating terms. The need to feed the actual output signal back into the modulator makes the direct generation of PWM from an SPDIF source unattractive .

Advantages

Despite the complexity involved, a properly designed class-D amplifier offers the following benefits:

* Reduction in size and weight of the amplifier,
* Reduced power waste as heat dissipation and hence smaller (or no) heatsinks,
* Reduction in cost due to smaller heat sink and compact circuitry,
* Very high power conversion efficiency, usually ≥ 90%.

This still leaves the signal with significant out-of-band content, which may be filtered out. To maintain a high efficiency, the filtering is done with purely reactive components (inductors and capacitors), which store the excess energy until it is needed instead of converting it into heat.

Table of Contents

1.0 Abstract …………………………………………………………….. 2

2.0 Introduction ………………………………………………………….3

3.0 Circuit Components ……………………………………………………4

3.1 Triangle Wave Generator ………………………………….….4

3.2 Pulse Width Modulator ……………………………….……….6

3.3 Tone control filters ………………………………………..……8

3.4 Volume and balance control filters …………………………..10

3.5 H-Bridge amplification stage ………………………… ……..11

3.6 Demodulation filter ………………………….………………..12

4.0 Conclusion …………………………………………………………….14

1.0 Abstract

This project consisted of the design and construction of a two channel, 10Watt class D audio amplifier with a carrier frequency of 44kHz, along with volume, balance, and tone controls. The main reason for the use of class D amplifiers is there extreme power efficiency. Class D amplifiers are composed of a few essential components. First the audio signal is converted into a pulse train using a pulse width modulator. That signal is then sent through a switched mode power gain stage and then the signal is demodulated with a low pass filter. The results of the project showed that the sound quality was relatively good for the carrier frequency specified above. However the tone controls were not as successful as we had hoped for especially the treble (high frequency) control.

2.0 Introduction

Class D amplifiers use pulse width modulation techniques to achieve a very power efficient amplifier. Class D amplifiers use transistors that are either on or off, and almost never in-between, so they waste the least amount of power. Class B amplifiers use linear regulating transistors to modulate output current and voltage and they can never be more efficient than 71%. Obviously, then, class D amplifiers are more efficient than class A, class AB, or class B. Some class D amplifiers have greater than 80% efficiency at full power. Class D amplifiers can also have low distortion, although not as good as class AB or class A. Because of the high power efficiency, they are ideal to use in small or portable electronics because they do not require large heat sinks to cool the transistors. The general components that make up the class D amplifier are a pulse width modulator, a switched mode power gain stage, and a demodulation filter.

Class D amplifiers have been around since the 60’s but were never very successful for audio applications because they had such high distortion. With the invention of the power MOS transistor, class D amplifiers suddenly became useful because the MOS transistor allowed for a very fast switching frequency with little distortion. Even with the Power MOS transistor the distortion level for high frequency signal can still be substantial. Today they are best used for subwoofers or low frequency signals in audio applications.

3.0 Circuit Components
The amplifier was composed of only a few components shown in the block diagram below.

All of the components up until the PWM will be referred to as the pre-amp in later sections. Over the course of the descriptions and explanations to come please refer to the complete attached schematics of the entire circuit.

3.1 Triangle Wave Generator
A triangle wave was needed to convert the audio signal into a pulse width modulated signal using a comparator. The triangle wave generator that we made consisted of an integrator and a hysteresis comparator. From an intuitive perspective, all the circuit did was integrate a square wave that was created by the hysteresis comparator. The biasing for the input and the positive feedback resistors of the hysteresis comparator were chosen such that the output would switch to the opposite rail when the input was at +/- 10 volts. This was done by setting the feed back resistor (R18) to 1k ohm, and analyzing the voltage divider network between the nodes A and B. The voltage at node B is limited by

the diodes to +/- 0.9 volts. By setting the voltage at the positive input to the op-amp to
zero(the point of witching) the value of R16 needed to switch the comparator at an input voltage of +/- 10 volts was found to be 8.5k ohms. When the comparator flips to the opposite rail, that signal then feeds around to the integrator, and the integrator begins to integrate the same function but of opposite sign from before. Thus a triangle wave comes out of the output of the op-amp at node A. If you refer to the calculations section(5.0), greater input resistance results in a smaller slope of the output waveform and a smaller input resistance results in a greater slope. Therefore adjusting the pot(R14) changes the frequency of the triangle wave by dictating how fast the output of the integrator climbs until it hits the voltage at which the comparator switches.

3.2 Pulse Width Modulator:
The pulse width modulator took the audio signal and converted it into a pulse signal with varying duty cycle that was proportional to the input signal at each sampled point. This was accomplished by sending the triangle wave and the audio signal from the pre-amp filters through a comparator. The results are shown below. As can be seen from the

oscilloscope plot of the waveforms, the triangle wave samples the input audio signal at every point that the two curves cross and converts it into a series of pulse signals of various lengths. A pull-up resistor was needed on the output of the comparator because it had an open collector output. Without the pull-up resistor the output square wave had a peak voltage of 500mv, but with the resistor the peak voltage was 16v. Converting the audio to a PWM signal is desirable because it is better to use a square wave to drive the power gain stage because there is far less power dissipated when the transistors are turned on and off by a square wave rather than turning them on slowly. In an effort to reduce some of the noise and oscillations in our amplifier, we put bypass capacitors around the positive and negative supply pins on the comparator. It helped somewhat in reducing the noise in the circuit.

One problem that was important to avoid, and one of which we learned the hard way was over modulation in the PWM. This occurred when the amplitude of the input signal exceeded the amplitude of the triangle wave. When this occurred the output signal became very distorted because there are point where the triangle wave goes through a whole cycle without intersecting the audio signal.

3.3 Tone Controls
A schematic of the tone controls is shown below. The filter on the left is the filter for the low frequencies and the filter on the right is the filter for the high frequencies. The low

frequency filter is an inverting op-amp configuration with a pot that changes the feedback and input resistance. When the pot is turned all the way to the right the input resistance is large and the gain is small and when it is turned all the way to the left the feedback resistance is large and the gain is large. The capacitors are there to nullify the affect of the pot at the frequency at which they act as a short. When that frequency is reached the position of the pot is irrelevant and the feedback and input resistances are the same, which yields a gain of one for all frequencies above the desired frequency. The frequency chosen was 1kHz and can be seen in the bode plot above. So by adjusting the pot the gain is varied between +/- 20dB for all frequencies up to 1kHz. So in reality this acted as the bass control for the audio signal.

The high frequency filter operated in a similar but inverted fashion. Unlike the low frequency filter where the gain was unity for all frequencies past 1kHz, the gain for the high frequency filter is unity for all frequencies up to 1kHz. When low frequency signals enter the filter the capacitors are open and therefore the setting on the pot has no affect on either the feedback or input resistances. This results in a gain of one because the feedback and input resistances are equivalent. When signals of frequencies greater than 1kHz enter the filter the pot adjust the gain of those signals by changing the ratio of feedback resistance to input resistance. This high frequency filter was the treble control for the audio signal. It was important to make the filters such that they did not over lap in frequency because the two filters were connected in series. When two things are

connected in series the overall gain is the product of the gain from each individual stage. So any signal that got amplified from one filter had a gain of unity in the other filter, so that no signal would be amplified twice. The overall range of the two filters in series can be seen in the bode plot above. The calculations for the component values can be found in section 5.0.

3.4 Volume and Balance
I will not elaborate much on the volume and balance controls for the amplifier because they were quite simple.

Both channels of the audio signal (left and right speakers) went through identical volume controls. The value of the input resistance for the volume control was chosen such that with the pot turned to 100% the volume of the music was at the desired maximum level.

The balance control was very similar. For this we wanted a gain of unity when the pot was turned to 100%. To make the two channels work opposite to each other we used a tandem pot with the feed back loop for the second channel connected to the opposite end of the pot from the first channel. So when the pot was at 100% one channel had unity gain while the other channel had a gain of zero. When the pot was at 50% the gain for both of the channels was equivalent.
3.5 H-Bridge
An H-bridge was used for the power amplification of the PWM signal. Shown in the schematic below, an H-bridge is a rectangular arrangement of transistors with a load

across the center. The idea is to drive the H-Bridge with a square wave on each side of bridge, with the driving signal on one side one half cycle out of phase from the other side. In our circuit we used two p-channel MOSFET’s for the two top transistors and three n-channel MOSFET’s, two for the bottom and one to invert the driving signal for the other side of the bridge. We chose to use two p-channel MOSFET’s for the top portion of the bridge because we did not want to use a separate IC or have a big messy thing of circuitry to drive a bridge made of all n-channels. P-channels turn on in exactly the opposite manner than that of an n-channel so it made sense to drive both the gates by the same signal. In order to make the H-Bridge work, we needed to turn on the MOSFET’s in diagonal pairs. This allowed a path of current to flow from +Vcc to ground across the load but twice as much voltage swing because the current is in the opposite direction across the load when the other diagonal pair turns on. By inverting the driving signal on
the right side and using that to drive the left side of the bridge, the bridge began to function properly. The simulations showed a fairly clean amplified signal across the load resistor with amplitude of about 8.5volts and an output power of about 12 watts. The output power for the H-Bridge is directly proportional to the variation in duty cycle in the PWM signal. As it was discussed earlier, when the triangle wave is much greater than the audio signal that it is sampling the variation of the duty cycle in the PWM signal is

very small, therefor the average power dissipated across the load in the H-Bridge is very small. When the amplitude of the input audio nears the amplitude of the triangle wave the output power becomes much louder because the variation in the duty cycle is much greater.

3.6 Demodulation Filter
The demodulation filter that we attempted to use was a double pole roll off LC filter. We used this to filter the PWM signal after it has gone through the H-Bridge. The pole was

placed at 20 kHz because that is the only portion of the signal that we really care about because the human ear can only hear up to that frequency. We chose an LC filter because it has a roll off that is twice as fast as would a first order low pass RC filter. In our actual project, we were unable to find inductor sufficient enough to meet our needs so the Filters were omitted due to time constraints.

4.0 Conclusion

Overall the project was a success. This project provided a sound medium in which I could enhance my knowledge and experience in electronics. Prior to embarking on this project I knew nothing about class D amps, pulse width modulation, active filters, and H-Bridges. Having spent this quarter researching and studying these things I feel I have learned a lot and have broadened my interests and made available new subjects of interest that I was previously unaware of. Perhaps one the most important skills that I really developed over the quarter was trouble shooting. It is very easy to get frustrated when things don’t work out like they should, especially when it comes to soldering the final product. Fortunately the four errors that we made in soldering up the circuit were ones that we discover and fixed in a relatively short amount of time.

One aspect that still has me somewhat puzzled was the fact that we were never able to get a really clean signal from our output. We tried putting bypass capacitors around the +/- Vcc to the comparator so that it would switch with the least amount of noise in the output wave and that seemed to give some improvement but it was not substantial. All in all I think that this project has bettered my understanding of electrical engineering knowledge and also time management skills.